Low-loss tunable ferro-electric device and method of characterization

ABSTRACT

A tunable ferroelectric component and a narrowband resonant circuit for measuring the loss of the ferroelectric component. The ferroelectric component may be a capacitor integrated in the resonant circuit. The testing method eliminates other sources of loss to isolate the loss due to the ferroelectric material and to demonstrate that this loss is low.

RELATED APPLICATIONS

[0001] This application claims the benefit U.S. Provisional Application60/283,093, filed Apr. 11, 2001, which is hereby incorporated byreference. Additionally, this application relates to U.S. application“Tunable Ferro-electric Filter,” filed on Jul. 13, 2001, and U.S.application “Tunable Ferro-electric Multiplexer,” filed on Jul. 24,2001, which are hereby incorporated by reference.

FIELD OF THE INVENTION

[0002] The field of the present invention is ferro-electric tunableelectronic devices and components.

BACKGROUND OF THE INVENTION

[0003] Variable capacitors are advantageous as different electronicresponses can be obtained by variation of the capacitance. Thestructures presently used to implement variable or tunable capacitors,however, have significant performance and practical limitations. Movableparallel plates, while providing variable capacitance for radio tuning,are bulky, lossy, noisy, generally operate over only a limited range offrequencies, or have any number of these limitations. A “lossy”component or device has a high insertion loss (IL), which is the ratioof power dissipated in the component to power delivered to a load. Anelectronic varactor is a semiconductor device that adjusts capacitanceresponsive to an applied voltage. Varactors are typically lossy andnoisy, and are therefore generally ineffective for high-frequencyapplications, particularly those above 200 MHz. Hence, they are notsuited for tuning insertion loss-critical devices such as filters andmultiplexers in wireless applications, particularly where Code DivisionMultiple Access (CDMA) is used. Another implementation providingvariable capacitance is a micro-electro-mechanical system (MEMS). Thisis a miniature switching device that physically selects a differentcapacitor responsive to an applied signal. MEMS, however, is typicallycostly, unreliable, requires a substantial control voltage, and enablesonly a discrete set of pre-selected capacitance values.

[0004] Because of their variable dielectric constant, ferroelectricmaterials are good candidates for making tunable capacitors or othertunable components. Under presently used measurement andcharacterization techniques, however, tunable ferroelectric componentshave gained the reputation of being consistently and substantiallylossy, regardless of the processing, doping or other fabricationtechniques used to improve their loss properties. They have thereforenot been widely used. Ferro-electric tunable components operating in RFor microwave regions are perceived as being particularly lossy. Thisobservation is supported by experience in RADAR applications where, forexample, high RF or microwave loss is the conventional rule for bulk(thickness greater than about 1.0 mm) f-e materials especially whenmaximum tuning is desired. In general, most f-e materials are lossyunless steps are taken to improve (reduce) their loss. Such stepsinclude, but are not limited to: (1) pre and post deposition annealingor both to compensate for O₂ vacancies, (2) use of buffer layers toreduce surfaces stresses, (3) alloying or buffering with other materialsand (4) selective doping.

[0005] As demand for limited range tuning of lower power components hasincreased in recent years, the interest in ferroelectric materials hasturned to the use of thin film rather than bulk materials. Theassumption of high ferroelectric loss, however, has carried over intothin film work as well. Conventional broadband measurement techniqueshave bolstered the assumption that tunable ferroelectric components,whether bulk or thin film, have substantial loss.

[0006] A broadband measurement of the capacitance value of aferroelectric capacitor is typically obtained using a device such as anLRC meter, impedance analyzer or a network analyzer. From powermeasurements, one can calculate the lossiness of the capacitor. Theinverse of lossiness is referred to as the Quality Factor (“Q”) Thus, alossy device will have a low Q and a more efficient device will have ahigh Q. Q measurements for ferroelectric capacitors with capacitances inthe range of about 0.5 pF to 1.0 pF operating in a frequency range of1.8 GHz to 2.0 GHz, obtained using conventional measurement techniques,are typically claimed to be in the range of 10-50. This is unacceptablyinefficient, and ferroelectric tunable components are thereforeconsidered undesirable for widespread use. In wireless communications,for example, a Q of greater than 80, and preferably greater than 180,and more preferably greater than 350, is necessary at frequencies ofabout 2 GHz.

[0007] As will be shown below, conventional ferroelectric componentshave been wrongly fabricated, measured and characterized. As a result,it is commonly assumed that ferroelectric tunable components are verylossy with Qs in the range of 10-50 in the L-band. Ferroelectric tunabledevices operating in other frequency bands have also been labeled ashaving Qs unacceptable for most applications.

SUMMARY OF THE INVENTION

[0008] The methods of testing the loss, or its inverse, Q, of f-e filmsare flawed in the prior art. The prior art methods typically usebroadband testing methods and non-integrated components. All of the lossmechanisms of the testing methods and devices under test are typicallynot accounted for. This has led investigators to believe that f-ematerials are lossy.

[0009] The invention provides for narrowband testing methods andintegration of components. All loss mechanisms are accounted for andeliminated or minimized. This results in more accurate test results,showing that some f-e materials are much less lossy than previouslythought.

[0010] With this testing method, f-e materials can be successfullyinvestigated to find ideal tradeoffs between loss, tunability and otherparameters. Low loss tunable f-e devices can be built. A low loss, f-etunable capacitor can be built. Such a capacitor can be used as abuilding block in many applications where tunability is desired, but lowloss requirements had been preventing the use of any other tunabledevices. An example application is in wireless communication devices.

BRIEF DESCRIPTION OF THE DRAWINGS

[0011] The details of the present invention, both as to its structureand operation, may be gleaned in part by study of the accompanyingdrawings, in which like reference numerals refer to like parts, and inwhich:

[0012]FIG. 1 is a top view of an interdigital capacitor fabricated witha thin ferroelectric film;

[0013]FIG. 2 is a sectional view of a tunable ferroelectric gapcapacitor according to the present invention;

[0014]FIG. 3 is a table showing the relationship between gap width,ferroelectric layer thickness and capacitance;

[0015]FIG. 4 is a top view of a ferroelectric overlay capacitoraccording to the present invention.

[0016]FIG. 5 is an exploded view of a portion of the overlay capacitorof FIG. 4.

[0017]FIG. 6 is a second order narrowband resonant testing circuitaccording to the present invention;

[0018]FIG. 7 is a table comparing data obtained with the testing circuitof FIG. 6 with data obtained using conventional test methods;

[0019]FIG. 8 is another embodiment of a second order narrowband resonanttesting circuit according to the present invention;

[0020]FIG. 9 is a single resonator narrowband testing circuit accordingto the present invention;

[0021]FIG. 10a is a schematic of another embodiment of a singleresonator narrowband testing circuit according to the present invention;and

[0022]FIG. 10b is a planar realization of the circuit schematic of FIG.10a.

DETAILED DESCRIPTION OF THE INVENTION

[0023] The present invention provides test methods utilizing narrowbandresonant circuits that accurately measure and characterize theefficiency of tunable ferroelectric components in the frequency range inwhich they will be used, and with topologies that will be used. Thesetest methods and circuits establish that tunable ferroelectriccomponents are not as uniformly lossy as previously thought and thatthey may advantageously be used in low-loss applications and devices,such as wireless handsets. With an accurate determination of loss,tunable ferroelectric components can be properly optimized and designed.Specific loss mechanisms can be identified and eliminated, or otherwisereduced and bounded.

[0024] Tunable ferroelectric components, especially those using thinfilms, can be employed in a wide variety of frequency agile circuits.Tunable components are desirable because they can provide smallercomponent size and height, lower insertion loss or better rejection forthe same insertion loss, lower cost and the ability to tune over morethan one frequency band. The ability of a tunable component that cancover multiple bands potentially reduces the number of necessarycomponents, such as switches that would be necessary to select betweendiscrete bands were multiple fixed frequency components used. Theseadvantages are particularly important in wireless handset design, wherethe need for increased functionality and lower cost and size areseemingly contradictory requirements. In CDMA handsets, for example,performance of individual components is highly stressed. Ferroelectricmaterials may also permit integration of RF components that to-date haveresisted shrinkage, such as an antenna interface unit (AIU) for awireless device.

[0025] For example, an AIU could integrate one or more tunable duplexers(US PCS and cellular in a dual band wireless communication device),diplexers, PA's and LNA's. Some or all of these components could beadvantageously integrated, their total size or volume or both beingreduced and their electronic performance improved. Further applicationsfor tunable ferroelectric components are set forth in the latter portionof this specification.

[0026] As with any dielectric, ferroelectric material has two primaryloss mechanisms, conductivity loss and damping from lattice vibrationsin the dielectric. The combination of the two effects is referred to asthe material's loss tangent (tan(δ)). For ferroelectric materialsconsidered in tunable RF or microwave circuits, damping from latticevibrations dominate, as there are no free charge carriers. However, anymethod which measures tan(δ) will include effects of finite conductivityif present. This is because the loss effects of the two machanisms areindistinguishable as far as rf/microwave properties are concerned.

[0027] A primary component in RF circuits is the capacitor. F-Etunability will now be discussed in terms of f-e capacitors. The totalloss of a capacitor, whether tunable or not, is given by its qualityfactor (Q) which is expressed as a ratio of its stored to dissipatedenergy, where the energy is stored in the electric field and dissipatedin resistance. For a lumped element capacitor, the unloaded Q (Q_(u)) isgiven by:

Q _(u) =X/R _(s)=1/(ω*R _(s) *C)  (1)

[0028] where ω=radian frequency; R_(s)=the series resistance of thecapacitor; and C=the capacitance of the capacitor. R_(s) is measured andgiven that C and ω are known, Q_(u) can be calculated. The seriesresistance arises from both the conductor and dissipative loss in thedielectric, i.e. tan(δ).

[0029] If a tunable capacitor is integrated into a resonant circuit, thetotal Q (Q_(t)) of the system is now given by:

1/Q _(t)=1/Q _(c)+1/Q _(d)+1/Q _(r)  (2)

[0030] where Q_(c) is the conductor Q; Q_(d) is the dielectric Q andQ_(r) is the radiation Q. For a well designed non-radiating system,there is no radiation loss. Hence, the conductor loss and the dielectricloss determine the total loss. The dielectric loss is the effect of theloss tangent, tan(δ), including conductivity loss attributable to thedielectric, if the latter loss is present. Hence, for both the unloadedQ and the total Q, a correct measurement of tan(δ) is crucial indetermining whether a tunable device can be fabricated with acceptableloss characteristics.

[0031] Cavity resonator methods are conventionally used to measure amaterial's dielectric constant and loss tangent. These methods aredifficult, especially at lower microwave frequencies (˜2 GHz) wherecellular phones operate, as the size of the cavity is quite large. Useof cavity resonator methods on thin ferroelectric films poses a greaterproblem, as it is very difficult to measure the perturbation introducedto a cavity from a structure having a thickness in the range of onemicron. The potential for error is significant.

[0032] Because of this difficulty with resonator methods, interdigitalcapacitors (IDC's) are usually used to measure ferroelectric filmquality. A ferroelectric interdigital capacitor (IDC) 100 in aconventional microstrip configuration is depicted in FIG. 1.Interdigital capacitor 100 comprises base substrate 110; thin filmferroelectric layer 120; and first and second conductors 130 and 140.Interdigital capacitors are typically used in applications such asmonolithic microwave integrated circuits (MMICs) and in applicationswhere small footprints and capacitances in the range of 0.1-6 pF areneeded. In an interdigital capacitor, the capacitance is created betweenconductive parallel lines or fingers in the structure.

[0033] Base substrate 110 typically comprises a low loss material suchas magnesium oxide (MgO), sapphire or high purity aluminum, for example.The substrate is chosen based on its inherent low loss tangent and itsability to accept the direct deposition of a wide range of f-e filmswithout additional buffer layers. A thin ferroelectric film 120 isdeposited on base substrate 110. Ferroelectric film 120 typically has athickness in the range of 0.15-1.5 microns. A conductive layer is thendeposited onto ferroelectric film 120. Sometimes an adhesion layer isneeded. The conductive layer is preferably a metallic material such ascopper, gold or silver. These metals are advantageous due to theirrelatively low loss at room temperature. For purposes of thisspecification, room temperature is defined as being in the range from−30° C. to +85° C. which covers the typical operating temperature rangefor most commercial components. The conductive layer typically has athickness in the range of 0.5 to 6.0 microns, with a thickness in therange of 0.5 to 1.5 microns being most common. Thickness requirementsvary based on skin depth which varies based on frequency.

[0034] While thin film (t_(f-e) less than about 1.5 μm) f-e materialshave been discussed, thick film f-e material can be used as well. Here,“thick film” is defined to be t_(f-e) greater than about 1.5 μm and lessthan about 1.0 mm. Bulk is greater than about 1.0 mm. The fabricationand application of thick film f-e material is quite different than thatof thin film f-e material. It usually involves a paste or a sol-geltechnique, and the f-e materials to produce the significantly addedthickness. The added thickness and especially reduced cost comes at theprice of somewhat degraded f-e performance, notably, reduced tunability.

[0035] Interdigital capacitor 100 is then fabricated using eitheretch-back or lift-off techniques to form first conductor 130 and secondconductor 140. First conductor 130 has fingers 132 and spaces 134 thatare proximate fingers 142 and spaces 144 of second conductor 140. Theconductors are arranged so that fingers 132 of first conductor 130 arein spaces 144 of second conductor 140, and so that fingers 142 of secondconductor 140 are in spaces 134 of first conductor 130. To date, mostresearchers and other practitioners in f-e film fabrication andcharacterization have designed IDC's with fingers typically 1-5 micronswide, and the gap or space between the fingers typically 1-5 micronswide.

[0036] The capacitance is created primarily between fingers 132 and 142.To generate a high level of capacitance, small gap size (<5 microns) andlong fingers are required. When used as a ferroelectric tuningcapacitor, small gap size also assists in creating a large tuning fieldbetween the fingers. This is critical because much of the tuning fieldis lost in the air region above capacitor 100.

[0037] The greatest loss component in this configuration is in the oddmode generated in the finger region. The coupling between the parallellines can be expressed in terms of an even mode and an odd mode. Theeven mode occurs when both lines are excited in phase (usually taken tobe zero), and the odd mode occurs when the lines are excited 180 degreesout of phase. In microstrip circuits, the velocities at which the evenand odd modes propagate are different. The loss further increases when athin conductive layer (less than 1.5 microns), narrow finger width andgap spacing (either or both less than 5 microns) and sharp corners areused.

[0038] The standard procedure for measuring thin ferroelectric film lossvia an interdigital capacitor is as follows. As described above,approximately 0.5 microns of ferroelectric film is deposited on a lowloss substrate such as magnesium oxide. Then, a conductive layer havinga thickness of 1 micron or less is deposited to permit fabrication of aninterdigital capacitor of the smallest possible size. Finger width andgap spacing are both typically in the 1 to 5 μm range. Etch-back orlift-back techniques are used to form narrow, long fingers with sharpcorners. The resulting interdigital capacitor is characterized using abroadband measurement tool such as an LRC meter or an impedance ornetwork analyzer with probe tips that contact the capacitor.

[0039] Using this procedure, capacitors in the range of 0.2 to 1.5 pFare obtained, with Q's in the range of 10-100 at an operating frequencyof anywhere from approximately 500 MHz to approximately 2 GHz istypically measured. This loss is typically attributed entirely to theferroelectric film. These Q values are considered quite low and,consequently, ferroelectric tunable components are commonly assumed tobe high loss and unacceptable for many uses. In wireless communications,for example, a Q of greater than 100 and preferably greater than 250 isnecessary at frequencies in the range of 2 GHz for f-e capacitors in thevicinity of 1.0 pF. As will be described below, however, conventionalfabrication and loss measurement techniques do not yield a reliableindication of the actual loss attributable to the ferroelectric film.

[0040] As indicated in Equation (1), capacitor loss (whether tunable ornot) is proportional to the series loss R_(s) at radio frequency(f>about 500 MHz) where the effect of the large parallel resistance thatshunts the capacitance is negligible. The capacitor does not care whatthe source of the series loss is, only that there is a source. Forexample, for a 1 pF ferroelectric tunable capacitor to have anacceptably low loss (Q_(u)=250) at 2 GHz, the series loss must be only0.32Ω. The series loss includes the total loss from all sources arisingfrom the capacitor's use. In order to minimize or eliminate the sourcesof series loss, one must first account for each loss mechanism that ispresent. This will permit a more accurate determination of the lossattributable specifically to the ferroelectric film.

[0041] For f-e devices, the total loss is governed by summing eachsource contribution as follows:

L _(t) =L _(geom) +L _(attach) +L _(metal) +L _(sub) +L _(rad) +L_(meas) +L _(f-e);

[0042] where L_(geom) in is derived from the topology of the capacitor,

[0043] L_(attach) is loss due to device attachment,

[0044] L_(metal) is the total metal loss,

[0045] L_(sub) is the base substrate loss (if present),

[0046] L_(rad) is the radiation loss, both desired and undesired,

[0047] L_(meas) is the total loss arising from measurement errors, and

[0048] L_(f-e) is the f-e loss tangent.

[0049] This loss allocation can first be used to obtain an accuratevalue of L_(f-e) (or f-e tan δ) at the desired operating frequency inthe manner in which the f-e capacitor will be used. To correctly deriveL_(f-e), one must eliminate or constrain all of the other losscontribution sources just described. For example, L_(geom) will varyaccording to topology, being best for an overlay capacitor, worse for agap capacitor, and much worse for an IDC capacitor. Although this losscan be reduced and controlled, it is inherent to a device. Consequently,the choice of topology for a given f-e capacitor will affect the bestpossible Q_(c) attainable from the f-e capacitor. Electromagnetic (EM)software can establish a baseline loss for a desired geometry, assuminga lossless f-e film. This baseline loss represents the best (lowest)loss for a given geometry.

[0050] In general, a gap capacitor is easiest to fabricate. An IDC isnext easiest, and an overlay capacitor is hardest of these three.Compared to an IDC, the gap capacitor will have a better Q but lowercapacitance per unit cross section (W in FIG. 1a). The IDC's capacitanceis greater due to the use of a number of fingers per unit cross section.For many communication filter applications, however, large capacitance(C≧4.0 pF) is not needed. Thus, a gap capacitor often can provideadequate capacitance. The inherently high value of κ for most f-e filmshelps provide relatively high capacitance per unit cross section, W,compared to a conventional gap capacitor.

[0051] L_(attach) arises from discrete device attachment techniques,including, for example, solder, silver paint, or wire bonding. Theseattachment losses may be large and unpredictable. The lowest losses areachieved by direct fabrication of the f-e capacitor to the resonator orother RF circuitry, thus minimizing if not eliminating this losscomponent.

[0052] The inherent loss of a stand-alone f-e capacitor is of littleconsequence. What is of much greater consequence is any added lossarising from the attachment of the f-e capacitor to a circuit. Even ifthe f-e capacitor were lossless, should a large loss connection be used,the overall effect is that of a lossy f-e device. For example, if aQ≧250 at 2.0 GHz is desired for a capacitance of 1.0 pF, then the totalseries resistance R_(s) must be ≦0.32 ohm. Any additional loss will thusfurther reduce the Q of this capacitor. That this additional loss isexternal to the actual capacitor is irrelevant. Even unavoidable lossmechanisms, such as those due to mounting, for example, lower theeffective Q of the capacitor from the perspective of its effect on thesystem.

[0053] For minimum added loss, the connection between the f-e capacitorand the resonator should provide the lowest added resistance. Thus, theelectric currents and charges associated with the f-e capacitor shouldsee a minimum added loss. Conventional bonding or mounting techniques,such as (but not limited to) soldering, wire bonding or silver paint orpaste do not provide for such a low loss, controllable bond.

[0054] The added, unpredictable loss arising from the use of suchbonding methods degrade the realized Q regardless of whether or not thef-e capacitor is being used for resonator tuning purposes orcharacterization of an f-e film. Thus, for best performance (lowestloss) the f-e capacitor structure should be directly fabricated onto orwith the resonator it is meant to tune or onto other essential RFcircuitry. Only by direct fabrication can there be a minimum losstransition for electromagnetic (EM) sources (currents) from the f-etuning elements to the resonator. The desirable effects of direct f-ecapacitor fabrication onto or with a resonator can be enhanced by thelack of sharp corners or transitions.

[0055] Factors for L_(metal) include the surface roughness (SR) of themetal, metal thickness as compared to skin depth, δs, and conductivity.SR may be effectively eliminated as a factor if SR is less thanaproximately 10 micro inches root mean square (rms) for operatingfrequencies in the L and S band (1-4 GHz). The metal thickness may bereduced as a factor if the thickness is 1.5δs or greater, or effectivelyeliminated if the thickness is ≧5δs. For electrode contacts, metalthickness (t_(m)) can be approximately 1.5δs. For the case ofelectromagnetic resonators, where a travelling or standing wave must besupported, i.e., where the metal in question extends for an appreciablefraction of a wavelength (about 10% or greater), the metal thicknessshould be closer to about 5δs or greater.

[0056] Conductivity is best for Au, Cu, or Ag. Thus, L_(metal) can bereduced and controlled, but not eliminated as a factor. Its effect,however, can be calculated by expressions well known to those skilled inthe art, or by using line calculator tools available in commonly usedcircuit simulators, such as Eagleware or Touchstone. Further, precisefabrication control can bound geometric variations in L_(metal).

[0057] The loss contribution represented by L_(sub) may be minimized bychoosing a low loss substrate with a loss tangent less than 0.001 andpreferably less than 0.0005 at the operating frequency of interest.Suitable materials include >99% pure alumina, a best current choice forloss/cost benefits. Sapphire or MgO are better than alumina in that theyhave lower loss tangents, but they are more expensive. All thesematerials will accept many f-e thin films without buffer layers and havea surface roughness that is acceptable with little or no furtherpolishing. Semiconductor substrates are poor choices because of theirrelatively high conductivity. In addition to the factors of losstangent, surface roughness and price, suitable substrates should not bebrittle, can be fabricated as larger area wafers, and can be easilymetallized without extensive pre-processing.

[0058] Separating out L_(sub) from the total loss of a compositesubstrate (f-e film plus substrate) can be achieved by using EM field orcircuit simulation software. For example, Sonnet, Momentum, or IE3D maybe used. Thus, L_(sub) can be reduced significantly and calculatedprecisely.

[0059] L_(rad) can be eliminated by proper shielding and design, and sois typically not a factor. It should be noted that a wide variety offilters, especially planar filters such as combline or hairpin, dependupon radiative coupling to achieve their desired performance. In thesecases, one should ensure that the unwanted, stray coupling is reduced,if not eliminated.

[0060] L_(meas) can add significantly to the circuit loss error becausesmall, added loss significantly reduces the measured Q of thedevice-under-test (DUT) or system thus obscuring the intrinsic Q of theDUT. The conventional method for measuring dielectric constant and losstangent in a material is the cavity perturbation technique, which iswell known to anyone skilled in the art. At L-band, however, the size ofthe cavity becomes quite large. When characterizing thin films (asopposed to bulk) with film thickness ≦1.5 μm, such as f-e films, theproblem becomes very difficult as measurement errors can be severe.Furthermore, one should characterize an f-e capacitor (or filter) in amanner most similar to how it will be used. Thus, the preferred way tocharacterize f-e compounds or films is by microstrip resonatortechniques.

[0061] For measurements on resonant circuits, a network analyzer is thepreferred choice. To minimize measurement loss and attain the mostaccurate measurement using a network analyzer, loss to DUT should becalibrated out, a full two port calibration of the analyzer should beperformed and averaging should be used for calibration and measurement.

[0062] Through minimization or elimination of the device attachment,substrate, radiation and measurement error loss components, the totalloss becomes:

L _(tot) =L _(geom) +L _(metal) +L _(f-e) +ΔL _(misc)  (4)

[0063] L_(tot) is the total loss for a given ferroelectric capacitorgeometry, and L_(geom) and L_(metal) are integral parts of thatgeometry. Their presence is appropriate for determining the actual lossof a specific device, but they can be quantified and removed in order todetermine the loss due solely to the ferroelectric material. L_(geom)can be determined from an accurate electromagnetic simulation of thecircuit assuming a lossless ferroelectric material; and L_(metal) can bedetermined using the expressions for metal loss assuming conductivity,surface roughness (if applicable) and skin depth. ΔL_(misc) represents acombination of incomplete removal of the other loss mechanisms with thefinite bounds on L_(geom) and L_(metal).

[0064] This two-step process of (a) accounting for all loss mechanisms;and (b) eliminating or bounding these losses not only permits anaccurate determination of the ferroelectric loss, it also helpsestablish correct design guidelines for low loss tunable components.Correct knowledge of L_(f-e) allows one to first determine whether ornot the film under consideration can be used for a proposed application.Knowledge of L_(f-e) further provides a necessary baseline for any typeof optimum design using ferroelectric films. This knowledge is necessaryif one is to effectively trade-off loss tangent for tunability. Inshort, accurate fabrication and measurement techniques result inconsistent ferroelectric film loss characterization.

[0065] Based on this loss analysis, low loss tunable ferroelectriccomponents, and in particular tunable ferroelectric capacitors, can bedesigned, tested and implemented in a wide variety of applications.Design procedure and implementation based on this loss analysis forthree common types of capacitors—(1) gap capacitors, (2) overlaycapacitors and (3) interdigital capacitors—will now be discussed.

[0066] A ferroelectric tunable gap capacitor 200 is illustrated in FIG.2. Gap capacitor 200 comprises substrate layer 202; ferroelectric layer204 and metal layer 206 defining capacitance-inducing gap 208. Thefollowing design implementation minimizes losses from other sources andpermits an accurate determination of the loss due to the ferroelectricfilm 204. It assumes an operating frequency in the L-band (1-2 GHz) forwireless handsets, though the same methods could be applied in otherbands.

[0067] In one implementation, substrate 202 is a layer of 99.5% purealumina having a thickness in the range of 20-40 mils. Surface roughnessshould be less than or equal to about 5 μinch rms. Ferroelectric layer204 is a film of barium strontium titanate, Ba_(x)Sr_(1−x)TiO₃, (BSTO)having a thickness in the range of 0.15 to 2.0 microns. Using a filmthickness >1.0 μm maximizes capacitance and tuning range.

[0068] Adjusting the Ba/Sr fraction, doping or annealing are preferablychosen to provide the minimum tan δ while providing the required tuningrange. In one embodiment, x=0.5 (in Ba_(x)Sr_(1−x)TiO₃) for roomtemperature operation. Alternative ferroelectric materials could also beused. Metal layer 206 has a thickness of approximately 2.5 μm, whichmakes it suitable for electrode application. Gap 208 is 30-80 mils wide,and the edges should be rounded to maximize loss reduction. Thecapacitance demonstrated by gap 208 is in the range of 0.6 pF to 1.5 pFat 0 volts DC bias.

[0069] EM simulations indicate that for a capacitance of approximatelyone pF at two GHz, a gap capacitor has Q>700, assuming a loss tangent of0.002, or Q>300, assuming a loss tangent of 0.005. FIG. 3 is a tableshowing the relationship between gap width, ferroelectric layerthickness and capacitance. This data is very useful for target design ofgap capacitor test circuits. The results in FIG. 3 assume a 0.5 micronthick ferroelectric film with a dielectric constant of 1000 at 0V DCbias, a 40 mil thick substrate layer of 99.5% pure alumina, and a losstangent of 0.002 for the f-e film.

[0070] A ferroelectric overlay capacitor 300 according to the presentinvention is illustrated in FIG. 4. Capacitor 300 comprises substrate310; bias pad layer 320; ferroelectric layer 330; and capacitor padlayer 340. Bias pad layer 320 defines a DC bias pad and capacitor pad340 defines capacitor pad 342 and DC blocking capacitor pad 344.

[0071] In one implementation, base substrate 310 is alumina having athickness in the range of 20-40 mils. Bias pad layer 320 comprises abase electrode layer of silver having a thickness of approximately 2.0microns covered by a layer of platinum having a thickness ofapproximately 100 nm. The platinum layer prevents the silver layer fromoxidizing during growth of the ferroelectric layer. Layer 320 has a padbuilt-in for connecting a resistance in the range of 0.5 to 1.0 MΩ toprovide DC bias. If needed, a thin (10 nm) chromium layer may beinterposed between the alumina and silver to provide better adhesion.Ferroelectric layer 330 is a thin film of BSTO having a thickness ofapproximately one micron. Capacitor pad 342 has a minimum area of 8.0 by4.0 mils and is topped by electrodes of gold or silver that have an areaof approximately 4.0 by 4.0 mils. The DC blocking capacitor has acapacitance of at least 150-200 pF and an area of approximately 100 by100 microns. The total area of contact pad 344 is a minimum of 7.0 by8.0 mils.

[0072] An overlay capacitor has a minimum capacitance in the range of0.8-1.5 pF. As can be seen in FIG. 5, which is an enlargement of aportion of capacitor 300, the overlap area 350 of capacitor 300 is verysmall. In one implementation, overlap area 350 has a size of 0.3 mil by0.3 mil. This is based on a BSTO dielectric constant of about 1000 at 0volts DC and a film thickness of about 1.0 microns. The pads 342 and 320taper to and from capacitor overlay area 350. The taper is from 4.0 milsto about 0.25 mils in 1.0 mil distance.

[0073] The loss target for capacitor 300 is a Q of at least 350 at 2.0GHz for 1.0 pF. If needed, the ferroelectric film 330 can be furtheroptimized via doping, annealing or use of a buffer layer or layers.Finally, the change in capacitance should preferably be 2:1 (50%) orgreater for a change in bias voltage of 0-2.5 volts.

[0074] One aspect of the present invention is optimal structures anddesign criteria for tunable ferroelectric components, of which thecapacitor structures described above are one example. Another aspect ofthe present invention is measurement methods and apparatus foraccurately characterizing the losses in tunable ferroelectriccomponents. These methods involve the use of resonators and narrowbandresonant circuits. Narrowband measurements are appropriate since thedevices being measured are designed to operate at a narrowband offrequencies. Narrowband (resonant) measurements are also preferred asthe naturally enhance the effect of small losses making them easier tomeasure, and they make the measurement more accurate. Prior methods haveinvolved broadband measurements that are inappropriate and inaccuratefor narrowband devices. Two inventive implementations of these testingresonant circuits will be described: second order narrowband bandpassfilters, and microstrip resonator circuits (halfwave or quarterwave).

[0075]FIG. 6 shows a resonant narrowband testing circuit 400 configuredto test two ferroelectric capacitors 410 and 412. It is a 2^(nd) orderplanar combline filter. Capacitors 410 and 412 are configured asdescribed with respect to FIG. 1 and FIG. 2 and are implemented tominimize loss components. Testing circuit 400 comprises a planar, secondorder combline bandpass filter and includes two resonators 402 and 404coupled in series with, respectively, ferroelectric capacitors 410 and412. A DC bias voltage is applied to capacitors 410 and 412. Capacitors410 and 412 may be fabricated and mounted for testing either as lumpedelements or by printing directly on the substrate. DC blockingcapacitors (capacitance equal to about 180 pF) are not shown. In alumped configuration, the capacitors are soldered or attached withsilver paint or paste. This permits use of a wide variety of devices,however, there is an increased and unpredictable loss due to thismounting method. In a printed configuration, the capacitors are printeddirectly on the substrate. Printing is advantageous in that no solderingor bonding is required and there is a lower loss due to the directfabrication. The type of substrates that may be used is limited,however, due to the presence of the ferroelectric film. DC blockingcapacitors are not shown.

[0076] The response is measured through input and output lines 406 and408 connected to a network analyzer. A measurement of the resonatorcenter frequency f₀ permits determination of the actual capacitor value(see equation (1) above), and the insertion loss at f₀ determines thecapacitor Q. After these measurements are obtained, a circuit simulationcan be used to obtain capacitance and Q values and the results compared.

[0077] In order to demonstrate the dramatic difference in test resultsobtained using the test method of the present invention relative toconventional test methods, reference is made to FIG. 7. The table inFIG. 7 presents measurement data obtained from ferroelectricinterdigital capacitor samples fabricated at the Naval ResearchLaboratory (NRL), Washington DC, under contract to Kyocera WirelessCorporation (KWC), the assignee of the present invention. Capacitanceand Q measurements taken from the interdigital capacitor samples at NRLusing conventional test methods (in this case, an HP 4291B ImpedanceAnalyzer and a Cascade Tech microwave probe) are compared tomeasurements taken from the same samples at KWC using the novel testmethods described above.

[0078] For purposes of this experiment, the interdigital capacitors werefabricated to have a capacitance in the range of 0.5-1.2 pF; a gapspacing of approximately 5.0 microns; a finger width of at least 150microns; a ferroelectric film thickness of approximately 0.5 microns; ametal thickness in the range of 1.5-2.5 microns; and a finger lengthless than or equal to 100 microns.

[0079] The KWC testing circuit is configured in like fashion as circuit400. It is a second order planar Chebychev bandpass filter configured toresonate at approximately 1800 MHz. The interdigital capacitor samples,lumped element capactitors, were “flip-chip” mounted and attached usingsilver paint. Bias was applied to correct for the fact that typicallyC1≠C2, where C1 and C2 are the two combline bandpass filter loadingcapacitors required for correct operation of the filter. While C1 isintended to be equal to C2, in practice C1=C2 is rarely achieved. Themore common condition of C1≠C2 significantly increases passbandinsertion loss (as far as Q determination is concerned) if notcorrected.

[0080] High Q ATC and AVX chip capacitors in the range of 0.6 to 0.8 pFwere used to establish a baseline passband insertion loss. The Q's forthese chip capacitors were in the range of 600-800 at the testfrequency. An Eagleware circuit simulator was used to determine actualcapacitance and Q's for the interdigital capacitors to give the sameresonant frequency and passband insertion loss as the measured data.

[0081] The data in FIG. 7 is essentially worst case Q data, as noattempt was made to remove (calibrate out) all possible loss components.One such loss component includes bonding (attachment) losses which aredifferent for each line and interdigital capacitor. Another is theresulting resonator length mismatch; microstrip gap open end effectsbelow the location of the capacitors; and losses arising from the basicinterdigital capacitor geometry. This being the case, the difference inQ values obtained using the present invention relative to conventionalmethods is even more striking. Further reduction or elimination of errorsources such as, for example, the direct fabrication of gap capacitorsusing an alumina or MgO substrate will only improve the Q data.

[0082] Use of a second order bandpass filter as the narrowband resonanttest circuit has several advantages. Capacitor data can be extracted atthe operating frequency. The topology is simple, repeatable and easilyfabricated. The measurements are simple and there is little added errorby virtue of taking the measurements. The results are easy to compare tothe simulated results. There are also several disadvantages that shouldbe noted. The potential for difference in capacitance values describedabove may show up in the measurement data as increased loss. A smalladjustment in one of the bias voltages, however, can compensate for thisdiscrepancy. Also, stray capacitance and coupling can effect the f₀ andQ values obtained. These effects can also be accounted for via the EMfield simulator. Unequal mounting of f-e capacitors results in slightdifferences in the two resonator electrical lengths, which directly addsto I.L. Misalignment of the f-e caps can also result in added loss,manifesting itself as lower Q.

[0083] Another embodiment of a second order narrowband resonant testingcircuit 450 is depicted in FIG. 8. Testing circuit 450 takes the form ofa coaxial resonator tunable filter although other resonators, such asmonoblock, stripline or microstrip can be used. Again, ferroelectriccapacitors 452 and 454 may be lumped or printed. Test circuit 450further comprises coaxial quarter wavelength resonators 462 and 464.Non-ferroelectric capacitor 470 (C2) is coupled between resonators 462and 464, and non-ferroelectric capacitors 472 and 474 (C1) are coupledon the outsides of the resonators. This basic structure is aconventional fix-tuned 2^(nd) order top capacitively coupled BPF.

[0084] The measurement technique using circuit 450 is as follows. BPFperformance is measured first without, then with f-e capacitors inplace. In the first instance, the resonator center frequency f₀₁ and theinsertion loss IL₁ of the filter is first measured without ferroelectriccapacitors. In the second instance, the resonator center frequency f₀₂and insertion loss IL₂ of the filter are measured with ferroelectriccapacitors 452 and 454. Notably, f₀₁ will always be greater than f₀₂ andIL₂ will always be greater than IL₁ as long as resonators 442 and 444are the same length. The capacitance C_(fe) can be determined fromf₀₁-f₀₂, and Q(C_(fe)) can be determined from IL₂-IL₁ by comparison withsimulations to great accuracy. The f-e capacitors need not be added tothe original circuit. Rather, the basic top capacitively coupled BPF canbe fabricated with no f-e capacitors, and a second BPF fabricated withf-e capacitors directly. This would lead to a minimum-added-loss tunabletest circuit as it allows for direct fabrication of f-e capacitors withthe circuit.

[0085] An alternative testing circuit would involve the use ofphysically shorter resonators 442 and 444 when used in conjunction withf-e capacitors. This would cause the BPF to resonate at or near the sameresonant frequency as the non f-e BPF. The same f-e capacitor Qextraction method would be used.

[0086] Second order test circuit 450 has several advantages over secondorder test circuit 400. Both circuit 400 and circuit 450 are inherentlynarrowband structures but the coaxial resonators 462 and 464 can have avery high Q, resulting in a very low insertion loss. Very little straycoupling is involved due to the inherent shielding. Also, as withcircuit 400, test circuit 450 is not only a test circuit but could beused as a bandpass filter in real world applications. However, circuit450 is a little harder to fabricate and test. Fixturing is critical andadding the ferroelectric capacitors results in extra losses due tomounting. This can be overcome via direct fabrication of theferroelectric capacitors on the same circuit used to realize C1 and C2,and then having an additional circuit without the ferroelectriccapacitors.

[0087] The testing circuit and method can be further simplified by usinga single resonator rather than two. This eliminates the problem ofcapacitor mismatch. The resulting circuit is more robust, easier tomodel and less prone to errors. Note that though the results shown inFIG. 7, are the results of tests on interdigital capacitors, gap oroverlay capacitors can be advantageously used, since they both can havehigher Q's than interdigital capacitors.

[0088] A testing circuit 500 comprising a gap coupled microstripresonator in its simplest form is depicted in FIG. 9. Circuit 500comprises a low loss substrate 502, and a microstrip resonator 504separated from input line 506 by a gap 508. A ferroelectric thin film isdeposited in gap 508 to create the ferroelectric gap capacitor. Hence,resonator 604 and the gap capacitor are fabricated as a single,integrated structure. Alternatively, a ferroelectric material can bedeposited underneath the resonator 504, creating a tunable resonator.

[0089] Substrate 502 should be a high quality, low loss substrate suchas magnesium oxide, alumina having a purity of greater than 99% andsapphire Substrate 502 should also have a low S.R. (less than 5.0μinch). Resonator 504 can be either a half wavelength (open circuit) orquarter wavelength (short circuit) resonator. A half wavelengthresonator is longer but easier to fabricate, while a quarter wavelengthresonator is shorter but requires via. The width of gap 508 is chosenfor near critical-coupling.

[0090] A network analyzer is preferably used for the capacitance and Qmeasurements. The model for gap capacitance and expression for metalloss are used to extract the Q of the dielectric, which is now acomposite of the Q of the base substrate and the Q of the ferroelectricthin film. Hence, the added loss over that of the base substraterepresents the loss of the ferroelectric film. Finally, proper analysisof the measured data, such as that outlined in “Data Reduction Methodfor Q Measurements of Strip-Line Resonators,” IEEE Transactions in MTT,S. Toncich and R. E. Collin, Vol. 40, No. 9, September 1992, pp.1833-1836, hereby incorporated by reference, is required to accuratelyextract the Q, or loss, of the capacitor under test.

[0091] It is useful now to compare the second order narrowband resonanttest methods and circuits described with reference to FIGS. 6-8 with thegap coupled single resonator test method and circuit described withreference to FIG. 9. The gap coupled single resonator is advantageous inthat is small, simple and very easy to fabricate. It also requires notuning for any possible mismatch of the input and output capacitors C1.However, it is more difficult to extract the ferroelectric loss tangentfrom the overall substrate and coupling capacitor loss. The second orderresonant circuits, on the other hand, can be actual devices in additionto being testing circuits. Moreover, it is very easy to compare themeasured data to either simulation data or data obtained usingnon-ferroelectric capacitors with high Qs. The drawbacks of the secondorder circuits are that they are larger, more complex circuits and moretuning of the ferroelectric capacitors may be required to obtain minimuminsertion loss.

[0092]FIGS. 10a and 10 b depict a preferred narrowband resonant testingcircuit 600. Circuit 600 takes the form of a single resonator bandpassfilter. Referring to FIG. 10a, which is a schematic of circuit 600,circuit 600 comprises ferroelectric capacitor 610 coupled to resonator620. Capacitors 630 and 640 (C1) are input and output capacitorsconnecting the resonators to the measurement instrument.

[0093]FIG. 10b is a planar realization of circuit 600. As can be seen,capacitor 610 and resonator 620 are fabricated as an integratedcomponent. Ferroelectric film 616 is deposited on low loss substrate602. Resonator 620 and conductive pad 612 are separated by gap 614 overferroelectric film 616 to define ferroelectric gap capacitor 610. A DCbias voltage is applied to pad 612 and may include a bias resistor 625.DC blocking capacitor 618 is connected between pad 612 and ground.Capacitors 630 and 640 are realized by conductive strips 632 and 642deposited on substrate 602 that are spaced from resonator 620 to form acapacitive gap.

[0094] In one implementation, substrate 602 is formed from 99.5% purealumina and has a thickness of approximately 40 mils and an SR ofapproximately 5.0 μinch. Ferroelectric film 616 has a thickness ofapproximately 1.0 μm and is deposited in the region of gap capacitor 610only. Microstrips 612 and 620 have a thickness of 4-6 μm and are spacedby approximately 10 μm to define gap 614. The length of resonator 620 isselected so that the overall structure (capacitor 610 and resonator 620)resonates in the desired frequency band. In one implementation,resonator 620 is a quarter wave resonator. Further fabrication cyclescan be used to fine tune the resonant frequency if a specific resonantfrequency is desired or required.

[0095] Resonator 620 may be configured as a microstrip, coaxial orstripline resonator. A planar microstrip configuration is preferred asit facilitates easier extraction of the capacitance and Q values fromcircuit 600. The use of an integrated component structure (i.e., aresonator having an integrated gap capacitor, such as resonator620/capacitor 610) is advantageous relative to the use of a separateresonator and a lumped element capacitor as the unpredictable and hardto measure losses and errors introduced by a lumped element capacitorare eliminated.

[0096] A testing method using a single resonator bandpass testingcircuit, such as circuit 600, proceeds as follows. First, a singleresonator bandpass filter test circuit having an integrated gapcapacitor is fabricated as described above. Precise thin filmfabrication and processing techniques should be used to ensure that thedesired geometry and properties are attained. Preferably, a techniqueshould be used with tolerances in the range of ±0.5 microns. Once thecircuit is fabricated, the center frequency f₀ and insertion loss IL₀are measured. Preferably, these measurements are obtained using anetwork analyzer calibrated by means of a full two port calibration andusing averaging.

[0097] Next, the same circuit is designed and analyzed on anelectromagnetic field simulation tool such as Sonnet, IE3D or Momentum.Initially, the simulation assumes no loss due to the ferroelectric film(i.e., a loss tangent of zero). The ferroelectric dielectric constant isthen adjusted in the gap region to give the same center frequency f₀ asmeasured in the test circuit. IL₀ is then calculated for theferroelectric gap capacitor alone. This value is then used in thesimulation to account for the loss component L_(metal) associated withthe metal.

[0098] Next, another circuit simulation is run, but this time using anon-zero loss tangent. In one implementation, a loss tangent of 0.003 isused and IL₀ is recalculated. This iterative process is continued untilthe measured insertion loss IL₀ from the test circuit is obtained,thereby yielding a very accurate approximation of the loss tangent forthe circuit, as well as the loss component L_(geom) due to theparticular structure being tested (in this case, a gap capacitor).

[0099] The baseline performance of the SR-BPF can be established byfabrication of the circuit with no f-e film. The resulting resonantfrequency will of course be higher as the loading capacitor 610 issmaller. This result will provide accurate information on overall shapeand frequency response of the SR-BPF.

[0100] Circuit 600 is not only an accurate mechanism for measuring theloss introduced by a ferroelectric gap capacitor, it is also a basicbuilding block for low loss tunable filters that may be implemented in awide range of applications, such as wireless handsets. Narrowbandresonant circuits configured as taught herein can be used to enhance theefficiency of, and add tunability to, many components of a typical RFtransceiver. Examples of RF components in which the present inventioncould be implemented include, but are not limited to, duplexers,isolators, matching circuits, power amplifiers, multiplexers, bandpassfilters and low noise amplifiers. With each element being tunable, itbecomes unnecessary to use multiple circuitry blocks to accommodatemulti-band modes. If necessary, the resonant circuits can be cascaded inan appropriate fashion to create desired filters and systems, vastlyimproving system performance while decreasing cost and size. Many of thecomponents of a typical wireless handset would benefit from tunability.

[0101] The description and drawings contained herein are particularembodiments of the invention and are representative of the subjectmatter broadly contemplated by the invention. However, the inventionencompasses other embodiments that will be obvious to those skilled inthe art. Accordingly, the scope of the invention is limited only by theappended claims.

1. A method for determining loss associated with a ferroelectric circuitcomponent comprising: fabricating a circuit comprising the ferroelectriccomponent; measuring an insertion loss due to the ferroelectriccomponent; determining components of the insertion loss that are due toother loss sources; and removing the components due to other losssources from the measured insertion loss to determine the lossassociated with the ferroelectric component.
 2. A method as claimed inclaim 1, wherein the ferroelectric component is a gap capacitor.
 3. Amethod as claimed in claim 2, wherein the circuit comprises anintegrated structure including a resonator integrated with the gapcapacitor.
 4. A method as claimed in claim 3, the integrated structurecomprises conductive strips deposited on a low loss substrate separatedby a gap, and a thin film of ferroelectric material underneath the gap.5. A method as claimed in claim 1, wherein the ferroelectric componentis selected from a group comprising an interdigital capacitor; a gapcapacitor and an overlay capacitor.
 6. A method as claimed in claim 1,wherein the circuit is a narrowband resonant circuit.
 7. A method asclaimed in claim 1, wherein the insertion loss due to the ferroelectriccomponent is measured using a network analyzer.
 8. A method as claimedin claim 1, wherein the components of the insertion loss that are due toother loss sources are determined using a circuit simulation tool.
 9. Amethod as claimed in claim 1, wherein the components of the insertionloss that are due to other loss sources are determined using aelectromagnetic field simulation tool.
 10. A method as claimed in claim1, wherein the ferroelectric component has a Q greater than
 100. 11. Amethod as claimed in claim 1, wherein the ferroelectric component has aQ greater than
 200. 12. A method for determining the loss associatedwith a ferroelectric capacitor comprising: fabricating a narrowbandresonant circuit that integrates the ferroelectric capacitor; measuringthe center frequency and insertion loss of the circuit with a networkanalyzer; analyzing the circuit on a circuit simulation tool todetermine the components of the insertion loss due to conductive metalcomponents of the resonant circuit and due to the geometry of theferroelectric capacitor; removing these components from the measuredinsertion loss to determine the loss due to the ferroelectric capacitor.13. A method as claimed in claim 12, wherein the narrowband resonantcircuit comprises a microstrip resonator having a gap to define thecapacitor.
 14. A tunable thin film ferroelectric device fabricated usinga method that isolates the loss due to the ferroelectric film.
 15. Atunable device as claimed in claim 14, wherein the device comprises aferroelectric capacitor and a resonator.
 16. A tunable device as claimedin claim 14, wherein the device comprises a planar, second ordercombline bandpass filter coupled to a lumped element, interdigitalcapacitor.
 17. A tunable device as claimed in claim 14, wherein thedevice comprises a microstrip resonator having an integrated gapcapacitor.
 18. A narrowband resonant circuit having an integratedferroelectric capacitor, the circuit being configured to permit accuratetesting of the loss associated with the capacitor and to facilitate itsuse as a building block in a tunable circuitry component.
 19. Anarrowband resonant circuit comprising a microstrip resonator having anintegrated gap capacitor, wherein the resonator comprises thin metalstrips separated by a gap on a low loss substrate, the gap capacitorcomprises a ferroelectric film deposited proximate the gap between thestrips.
 20. A narrowband resonant circuit as in claim 19, wherein thegap capacitor has a Q greater than about
 100. 20. A method of testing atunable ferroelectric component configured to operate in a tunablefrequency range comprising: designing a resonant circuit configured toresonate in the tunable frequency range; coupling the resonant circuitto the tunable ferroelectric component; measuring the loss in thetunable ferroelectric component using the tunable circuit; anddetermining non-ferroelectric sources of loss associated with theferroelectric component to demonstrate that the ferroelectric lossassociated with the component is acceptably low.
 21. A tunableferroelectric capacitor comprising: a first conducting surface; a secondconducting surface, the first and second conducting surfaces comprisinga capacitor; a ferroelectric material proximate the first and secondconducting surfaces; a variable voltage line coupled to theferroelectric material for changing a capacitance of the capacitor,responsive to a changing dielectric constant of the ferroelectricmaterial, responsive to a voltage applied to the variable voltage line;wherein a Q of the capacitor, when operated in a temperature rangebetween about −50 degrees Celsius and 100 degrees Celsius, is greaterthan about 80, in a frequency range between 0.25 GHz and 7.0 GHz.
 22. Atunable ferroelectric capacitor as in claim 21, wherein the qualityfactor, when operated in a temperature range between about −50 degreesCelsius and 100 degrees Celsius, is greater than about 80, in afrequency range between about 0.8 GHz and 7.0 GHz.
 23. A tunableferroelectric capacitor as in claim 21, wherein the quality factor, whenoperated in a temperature range between about −50 degrees Celsius and100 degrees Celsius, is greater than about 80, in a frequency rangebetween about 0.25 GHz and 2.5 GHz.
 24. A tunable ferroelectriccapacitor as in claim 21, wherein the quality factor, when operated in atemperature range between about −50 degrees Celsius and 100 degreesCelsius, is greater than about 80, in a frequency range between about0.8 GHz and 2.5 GHz.
 25. A tunable ferroelectric capacitor as in claim21, wherein the quality factor, when operated in a temperature rangebetween about −50 degrees Celsius and 100 degrees Celsius, is greaterthan about 180 in a frequency range between 0.25 GHz and 7.0 GHz.
 26. Atunable ferroelectric capacitor as in claim 21, wherein the qualityfactor, when operated in a temperature range between about −50 degreesCelsius and 100 degrees Celsius, is greater than about 180 in afrequency range between about 0.8 GHz and 2.5 GHz.
 27. A tunableferroelectric capacitor as in claim 21, wherein the quality factor, whenoperated in a temperature range between about −50 degrees Celsius and100 degrees Celsius, is greater than about 80 for a capacitance in arange between about 0.3 pF and 3.0 pF.
 28. A tunable ferroelectriccapacitor as in claim 21, wherein the quality factor, when operated in atemperature range between about −50 degrees Celsius and 100 degreesCelsius, is greater than about 80, for a capacitance in a range betweenabout 0.5 pF and 1.0 pF.
 29. A tunable ferroelectric capacitor as inclaim 21, wherein the quality factor, when operated in a temperaturerange between about −50 degrees Celsius and 100 degrees Celsius, isgreater than about 180 for a capacitance in a range between about 0.3 pFand 3.0 pF.
 30. A tunable ferroelectric capacitor as in claim 21,wherein the quality factor, when operated in a temperature range betweenabout −50 degrees Celsius and 100 degrees Celsius, is greater than about180 for a capacitance in a range between about 0.5 pF and 1.0 pF.
 31. Acapacitor as claimed in claim 21, wherein the capacitor has acapacitance of about 0.8 to 1.5 pF when zero voltage is applied to theferroelectric material.
 32. A capacitor as claimed in claim 21, whereinthe ferroelectric material comprises barium strontium titanate.
 33. Acapacitor as claimed in claim 21, wherein the ferroelectric materialcomprises a film having a thickness of approximately one micron.
 34. Acapacitor as claimed in claim 21, wherein the capacitor is a microstripgap capacitor.
 35. A capacitor as claimed in claim 26, wherein the firstconducting surface and the second conducting surface are separated by agap approximately 2.5 microns wide.
 36. A capacitor as claimed in claim21, wherein the conductors are metal strips having a thickness in therange of 2-3 microns.
 37. A capacitor as claimed in claim 21, whereinthe capacitor is an overlay capacitor.
 38. A capacitor as claimed inclaim 21, wherein the second conducting surface comprises either gold orsilver.
 39. A capacitor as claimed in claim 21 wherein: a first taper tothe ferroelectric capacitor from a ferroelectric capacitor bond padcomprises a contraction of the first conducting surface from about 4.0mils wide to about 0.1 mils wide over a distance of about 1.0 mils; anda second taper from the ferroelectric capacitor to a DC bias pad regioncomprises an expansion of the second conducting surface from about 0.1mils wide to about 4.0 mils wide over a distance of about 1.0 mils. 40.A tunable ferroelectric filter comprising: a first element having aninductance; a second element having a capacitance, the first and secondelements being electrically coupled in a filter configuration to producea characteristic frequency; a ferroelectric material positioned neareither the first element or the second element; and a control linecoupled to the ferroelectric material for varying a dielectric constantof the ferroelectric material and the characteristic frequency; whereina Q of the tunable ferro-electric filter is greater than about 100.